Presently, the design of commercial cordless telephone systems is based primarily on analog signal processing and transmission techniques. The use of digital techniques in other transmission systems have resulted in improved system performance due to a reduction in signal interference and noise achieved using digital techniques. It is, therefore, desirable to incorporate digital signal processing and digital transmission techniques in the next generation cordless telephones.
Such cordless telephone systems typically include a battery powered portable station (handset) and a base station. The base station is optimally connected to other telecommunication networks. Although the invention may be used in any digital transmission system, its use will be described herein for application in digital cordless telephone (DCT) systems.
Communication channels between the handsets and base stations in DCT systems may be set up using slotted ALOHA, a well known TDMA (time division multiple access) technique. The DCT system may communicate, for instance, using TDD (time division duplexing) for transferring information between the handsets and the base station. It is typical in such systems to operate in both burst and continuous modes. The burst mode is generally used to broadcast messages and to transmit control information, i,e. to set up a link between the base station and a particular handset. Once all of the control functions have been performed to set up a link, data, i.e. voice data, may be transmitted using a series of continuous bursts, referred to as the continuous mode.
A common form of digital communication employs a digital modulation technique known as Phase Shift Keying (PSK). In PSK, the phase of a carrier signal is switched between two or more values in response to binary data representing the information to be communicated. Where only two transmit phases are provided, each phase represents a single binary digit. For instance, the carrier signal can be switched so that its phase is 180.degree. in response to a binary "1" and switched to 0.degree. in response to a binary "0". This technique is known as phase reversed keying (PRK). The PRK waveform can be written as EQU .o slashed..sub.1 (t)=A sin(.omega..sub.c t) (1) EQU .o slashed..sub.2 (t)=-A sin(.omega..sub.c t) (2)
where .omega..sub.c is the angular frequency of the carrier and .o slashed..sub.1 and .o slashed..sub.2 are the phases of the PRK signal. The PRK waveform according to equations (1) and (2) is shown in FIG. 1.
To increase bandwidth efficiency (the number of bits transmitted per unit of time), a technique known as quadature PSK (QPSK) is used. In QPSK each transmit phase represents two bits of data thereby increasing the amount of data that can be transmitted over each phase interval. The advantage of QPSK modulation is that both the in-phase (I) and the quadature (Q) portions of the carrier signal can be modulated and combined to form the QPSK signal. For instance, FIG. 2a. shows an unmodulated phaser of the carrier signal. FIG. 2b and FIG. 2c show the modulated carrier of each the I and Q portions of the carrier signal respectively. The QPSK signal can be represented by: EQU .o slashed.1=A cos(.omega..sub.c t) (3) EQU .o slashed.2=-A sin (.omega..sub.c t) (4) EQU .o slashed.3=-A cos (.omega..sub.c t) (5) EQU .o slashed.4=A sin (.omega..sub.c t) (6)
The phaser diagram shown in FIG. 2d results from the combination of the I and Q portions of the carrier signal.
FIG. 3 is a block diagram of a prior art coherent QPSK demodulator. As shown, the QPSK carrier signal is received and filtered by bandpass filter 500. Filter 500 rejects undesirable adjacent channel interference and thermal noise. Typically, automatic gain control (AGC) 502 is utilized to adjust the energy level of the received signal. In a TDMA system, large burst-to-burst level differences arising from downlink fading due to atmospheric attenuation, distance and scattering can vary significantly. Thus AGC 502 detects the peak power of the received signal and provides feedback to the receiver so that the receiver's amplifier levels can be adjusted according to the strength of the received signal. Power divider 504 is provided to compensate for the power level difference in the carrier phase and bit timing recovery circuits 506 and 508 respectively.
The carrier phase recovery circuit 506 extracts the I and Q signal components from the received PSK signal. The 90.degree. hybrid circuit 510 is used to separate the I and Q signals. To this end, the I signal is mixed with the cos (.omega..sub.c t) and the Q signal is mixed with the sin(.omega..sub.c t) by mixers 514 and 512 respectively. Integrators 518 and 516 are used to detect the energy of the down converted signal over each time interval according to a well-known relationship: ##EQU1## where f(t) is the signal (i.e., ideally f(t)=.o slashed.1, .o slashed.2, .o slashed.3 or .o slashed.4 over the interval t1 to t2) and E is the energy of the signal over the time interval t1 to t2. Since there are two transmit phases for each of the I and Q signals and they are separated by a 180.degree. phase shift, the phase of the signals over a given time interval is either +E or -E as shown in FIG. 4.
FIG. 4 shows that when the detected energy of the in-phase signal is -E, the probability that the received signal corresponds to transmit phase .o slashed.2 (equation 4) is greatest and when the detected energy of the in-phase signal is +E the probability that the received signal corresponds to transmit phase .o slashed.4 (equation 6) is greatest. Similarly, FIG. 4 shows that when the detected energy of the quadature signal is -E, the probability that the received signal corresponds to .o slashed.3 (equation 5) is greatest and when the detected energy of the quadature signal is +E, then the probability that the received signal corresponds to .o slashed.1 (equation 3) is greatest.
Referring again to FIG. 3, I and Q decision circuits 520 and 522 determine the transmit phase of the received signal and reconstruct the transmit data, i.e. the binary data represented by the phase of the signal. The reconstructed binary data output from the decision circuits 520 and 522 is then combined into a single serial stream of binary data by the parallel-to-serial converter 524.
In most transmission systems, including DCT systems, communication between a receiving unit and a transmitting unit requires burst synchronization. Such synchronization is typically accomplished by providing the demodulated binary data to a correlator which detects a known pattern, such as a predefined preamble. Detection of the preamble or other known pattern allows the demodulator to synchronize its timing with the received PSK signal so that the demodulator can decode the received symbols.
It is well known to fine tune the demodulator's timing to the received symbols during operation in a continuous mode to optimize system performance and reduce error. Such fine tuning may be provided by the symbol timing recovery circuit 508 shown in FIG. 3. A typical symbol timing recovery circuit would determine within which time intervals the maximum amount of energy is received. Those intervals should correspond to the symbol intervals of the received signal. Thus the symbol timing recovery circuit 508 causes the decision circuits 520 and 522 to determine the phase of the received signal so that the decision corresponds to only a single symbol.
It has been found that the analog demodulator of FIG. 3 can be simplified by digitizing the integration and decision functions. FIG. 5 is a block diagram of such a digital demodulator.
After demodulating the received signal using mixer 530, the PSK signal is sampled at a frequency greater than twice the Nyquist frequency, where the Nyquist rate is the highest frequency of the down converted PSK signal. It has been found that by determining the zero-crossings of the signal with respect to time and referencing the zero-crossing to a reference transmit phase, the phase of the received signal can be determined. Waveform digitizer 532 samples the down converter signal represented in FIG. 5 generally at 536. The zero-crossing digital signal processor (DSP) 534 estimates the zero-crossings of the sampled waveform and then compares them to those of each of the possible transmit waveforms to determine the phase of the received signal.
However, this technique can become quite complicated due to the iterative curve fitting for trigometric functions which is necessary to determine the phase of the received signal. Furthermore, noise, intersymbol interference, and timing misalignment degrade the received signal so that only a best curve rather than an exact curve can be identified.
To avoid these problems, a phase progression digitizing technique has been suggested. This technique bypasses the waveform digitizer 532 and the complicated zero-crossing DSP 534 by directly digitizing the signal phase. This technique uses a counter to count each cycle of the received PSK signal, either on up-crossings or on down-crossings. A fixed sample rate is selected to be at least equal to the Nyquist rate of the modulation, i.e. at least twice the symbol rate. Optimally a number of cycles will occur between samples. The samples mark events, i.e., an up-crossing or down-crossing, in time. Thus the phase of the received signal is determined by comparing the time of the events occurring in each symbol period.
For example, consider the phase progression plot shown in FIG. 6. The phase progression plot plots the events as a function of time. The PSK signal is shown below the plot. The samples or events are enumerated as well as the time of each event. The curve fit for determining the phase of each symbol becomes a system of parallel lines where each line corresponds to one of the possible transmit phases. Using this technique all amplitude information is discarded and trigometric curve fitting can be avoided.
Unfortunately, this technique has several limitations as well. In particular, the sampling frequency in such a scheme is critically linked to the signal frequency in that a sample must occur on either upward zero-crossings or downward zero-crossings. Thus, whenever the signal undergoes frequency drift, which is well know to occur in communication systems, or frequency changes for other reasons, the sample rate will require constant adjustment to track such frequency changes.
Accordingly, a need still exists for a digital demodulator which can detect the phase of the received signal regardless of frequency changes and drift of the received PSK signal, which is relatively inexpensive and simple to implement.